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Updated: 8 Mar 2007
  
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CONTENTS.
INTRODUCTION.
The 5534 is a low-noise, low distortion bipolar opamp. The 5532 is a dual 5534 with compensation set internally for unity-gain stability.
The 5534 is something of a minority The major features are:
- The bipolar input devices give their best noise performance at low to medium impedances, in the range 100 - 1000 Ohms. In virtually all audio applications the 5532 will be quieter than the TL072.
- Bipolar input devices, with significant bias and offset currents.
- The common-mode range of the inputs is a healthy +/-13V, with no phase inversion problems if this is exceeded.
- The input pins are tied together with diodes for reverse-Vbe protection. This is normally unnoticeable in audio use, but it does mean the 5532 makes a poor comparator.
- Very low, though not quite zero, THD levels. THD still depends on how heavily the output is loaded.
- Higher power consumption: approx 4mA per opamp section. The DIL version runs perceptibly warm when quiescent on +/-15V rails.
- Requires proper rail decoupling near the chip.
Fig **. Typical effect of bias current on a 5532 stage. 500 nA through 10K gives an offset of 5 mV. This has negligible effect on headroom, but is quite enough to cause switches to click and pots to crackle.
The inputs are connected together with back-to-back diodes for reverse-voltage protection; and should not be forcibly pulled to different voltages. The 5532 is intended for linear operation, and using it as a comparator is not recommended.
As can be seen from Fig 5, the 5532 is almost distortion-free, even when driving the maximum 500 Ohm load. The internal circuitry of the 5532 has to my knowledge never been publically explained, but it appears to consist of nested Miller loops that permit high levels of internal negative feedback.
The 5532 is the dual of the 5534, and is much more commonly used than the single.
SPECS.
Here are the vital statistics: All typical values, for +/-15V supply rails.
| Supply voltage | +/-20V abs max |
| Output range | +/-13V typ (2K load) |
| CM range | +/-13V |
| en | 4 nV/rtHz typ 1 kHz |
| in | 0.7 pA/rtHz typ |
| Ibias | 500 nA typ |
| Slew rate: | 9 V/us |
| Supply current | 8 mA total |
| Unity gain stable | YES (5532) |
| Cost | 58p RS Jan 2001 |
| Fig 5Aaah... that's better. Distortion is almost absent from the 5532, though loading still makes a detectable difference. (tex5532.gif) |
5532 and 5534 type opamps require careful supply-decoupling if they are to remain stable; otherwise they appear to be subject to some sort of internal oscillation that degrades linearity without being visible as oscillation on a normal oscilloscope.
The essential requirement is that the +ve and -ve rails should be decoupled with a 100nF capacitor between them, at a distance of not more than 2 inches. It is NOT necessary, and often not desirable to have two capacitors going to ground; every capacitor between a supply rail and ground carries the risk of injecting rail noise into the ground. The main rail decouple electrolytics can be used to do the job for several 5532/4 packages nearby, and this cost saving is an important layout point. While it is not normally necessary to decouple each package individually, it is probably best to err on the side of caution by doing so.
DECONSTRUCTING THE 5532.
To the best of my knowledge, virtually nothing has been published about the internal operation of the 5532. This is surprising given its unique usefulness as a high-quality audio opamp. I believe the secret of the 5532's superb linearity is the used of nested negative feedback inside the circuit, in the form of traditional Miller compensation.
Fig 3 shows the only diagram of the internal circuitry that has been released. This has been in the public domain for at least twenty years, so I hope no-one is going to object to my impertinent comments on it. I have spent quite a lot of time on opamp internals, so I hope I have some idea of what I'm talking about.
The circuit initially looks like a confusing sea of transistors, and there is even a solitary JFET lurking in there, but it breaks down fairly easily. There are three voltage-gain stages, plus a unity-gain output stage to increase drive capability. This has current-sensing overload protection. There is also a fairly complex bias generator which establishes the operating currents in the various stages.
In Fig 3 I have coloured the main signal paths, to aid comprehension. Red and blue are the input differential signals. In all conventional opamps these two signals have to be summed to create a single output signal, and the point or stage at which this occurs is called the "phase-summing point". After this the signal path is green. Please note that component and node numbers are mine.
Q1,2 make up the input differential amplifier. They are protected against reverse- biasing by the diode-connected transistors across the input pins. Note there are no emitter degeneration resistors, which would linearise the input pair at the expense of degrading noise. Presumably high open-loop gain (cf the number of gain stages) means that the input pair is handling very small signal levels so distortion is not a problem..
Q3,Q4 make up the second differential amplifier; note emitter degeneration is now present. Phase summing occurs at the output of this stage at Node 2. C1 is the Miller capacitor around this stage. Q5,6,7 are a Wilson current-mirror which provides a driven current-source as the collector load of Q4. The function of C4 is somewhat obscure but it appears to balance C1 in some way.
The third voltage-amplifier stage is basically Q9 with split-collector transistor Q15 as its current-source load. Q8 increases the basic transconductance of the stage, and C3 is the Miller capacitor around it- note that this Miller loop does NOT include the output stage. Things are a bit more complicated here as it appears that Q9 is also the sink half of the Class-B output stage. Q14 looks very mysterious as it seems to be sending the output of the third stage back to the input; possibly it's some sort of clamp to ensure clean clipping, but to be honest I haven't a clue.
Q10 plus associated diode generates the bias for the class-B output stage, just as in a power amplifier.
The most interesting signal path is the semi-local Miller loop through C2, which encloses both the second and third voltage amplifiers; both of these have their own local Miller feedback, so there are two layers of internal feedback. This is probably the secret of the 5532's low distortion.
Q11 is the source of the output stage, and as mentioned above, Q9 appears to be the sink. Q12,Q13 are the overcurrent protection. When the voltage drop across the 15 Ohm resistor becomes too great, Q12 turns on and shunts base drive away from Q11. In the negative half-cycle, Q13 is turned on, which in turn activates Q17 to shunt drive away from Q8.
The biasing circuit is probably not worth dissecting in detail, but there is one interesting point. Bipolar bias circuits tend not to be self-starting; no current flows anywhere until some flows somewhere, so to speak. Relying on leakage currents for starting is not wise, so here the depletion-mode JFET provides a circuit element that is fully on until you bias it off, and can be relyed upon to conduct as the power rails come up from zero.
I can see no reason why this couldn't be built with discrete components, for deeper investigation. Some of the operating currents and component values would have to be guessed, but it could be done.
